Method for Determining the Rotor Position of a Synchronous Machine Operated in Field-Oriented Manner

ABSTRACT

A method for determining the rotor position of a synchronous machine operated in field-oriented manner, which has an effective inductivity that is dependent on the rotor position,
     the motor current being acquired, and the motor voltage being set with the aid of a pulse-width-modulation method,   a signal which is in synchrony with the pulse-width-modulation frequency being superimposed on the motor voltage value to be set,   values of the motor current being acquired in synchrony with the pulse-width modulation frequency,   a current component induced by the superimposed voltage signal   and a residual current component, i.e., fundamental wave component, being determined,   the current component induced by the superimposed voltage signal being used for determining an estimated rotor angle position, whose phase error in relation to the actual rotor angle position is reduced by means of a flux model,   the residual current component being supplied to a current controller.

FIELD OF THE INVENTION

The present invention relates to a method for determining the rotorposition of a synchronous motor operated in field-oriented manner.

BACKGROUND INFORMATION

German Published Patent Application No. 102 26 974 describes a methodfor determining the rotor position of a synchronous machine operated infield-oriented manner. Here, a sensorless position detection isimplemented in that a high-frequency signal is additionally superimposedon the pulse-width-modulated output voltage of the inverter, and thehigh-frequency component in the associated current signal is separatedfrom the fundamental wave component. The high-frequency component isused for determining the position angle of the rotor. The fundamentalwave component is forwarded to a current controller, as actual value.

As an alternative, the high-frequency components would also be separablewith the aid of a bandpass filter.

German Published Patent Application No. 10 2008 025 408 describes amethod for regulating a synchronous machine, in which a rotor fluxvector which is perpendicular to the induced voltage is formed and usedas feedback quantity, which, when vanishing drift is present, vanishesas well.

SUMMARY

Example embodiments of the present invention provide for improving therotor position determination and improving the control method forsynchronous machines.

Among features of example embodiments of the present invention in themethod for determining the rotor position of a synchronous machineoperated in field-oriented manner are that an effective inductivity isprovided, which is dependent on the rotor position,

the motor current is acquired, and the motor voltage is set with the aidof a pulse-width-modulation method,

a signal which is synchronous with the pulse-width-modulation frequencyis superimposed on the motor voltage value to be set,

values of the motor current are acquired in synchrony with thepulse-width modulation frequency,

a current component induced by the superimposed voltage signal,

and a residual current component, i.e., the fundamental wave component,are determined,

the current component induced by the superimposed voltage signal is usedto determine an estimated rotor angle position, whose phase error inrelation to the actual rotor angle position is reduced by means of aflux model,

the residual current component is supplied to a current controller.

This has the advantage that a synchronous machine is operable without asensor, employing an improved control method. In particular, a currentvalue extrapolated for a future instant, which may be used as inputvalue for a control structure, is able to be determined from theacquired measured current values, so that the actuating variablesdetermined for the next time interval are able to be determined fromthis extrapolated current value and take effect at this instantsimultaneously. This improves the control method for the synchronousmachine. Reduced effective dead time, in particular, comes about in theclosed current control circuit, which makes it possible to enhance thecontrol response, especially the achievable bandwidth.

In example embodiments, a number of measured motor current values areused for determining parameters provided in a first parameter-basedfunction assigned to the fundamental wave component, and in a secondparameter-based function assigned to the current component induced bythe superimposed voltage signal, the sum of the parameter-basedfunctions in particular being adapted to the acquired current valuesand/or being adapted as fit functions. This offers the advantage that aparameter-based function is used for fitting, i.e., adapting, thefundamental wave component, which function has at least one non-linearcomponent. This also improves the determination of the superimposedhigher-frequency signal to be separated, since deviations in adaptingthe fundamental wave component lead to falsifications in thedetermination of the amplitude of the current component induced by thesuperimposed voltage signal, and the fundamental wave current componentis able to be determined very precisely.

In example embodiments, the first parameter-based function is apolynomial of the nth order, the number being greater than n by one, inparticular. This has the advantage that the polynomial is able to bedetermined precisely.

In example embodiments, the second parameter-based function is atriangular signal and/or a function composed of linear segments, itsamplitude being a parameter, in particular, and its frequency being insynchrony with the pulse-width modulation frequency. This has theadvantage that the high-frequency component is mathematicallyrepresentable in excellent approximation and may be described by onlyone parameter.

In example embodiments, the number is four, or greater than four. Thishas the advantage that a few current measurements already suffice toexecute the present invention.

In example embodiments, once the parameters of the parameter-basedfunctions have been determined, a current value is extrapolated for aninstant at which the values of the actuating variable, especially theoutput voltage, are made to take effect, so that a synchronous currentvalue is used as input value for the control structure when determiningthe values of the actuating variable. In this context it is advantageousthat an extrapolation of the fundamental wave component is able to beperformed, and the value entered in the control structure is thusassigned to the same instant as the values of the actuating variable.

In example embodiments, the amplitude of the current component inducedby the superimposed voltage signal is determined from a linearcombination of at least three of the detected current measured values.This has the advantage that the amplitude is able to be determined in avery simple manner.

In example embodiments, an extrapolated current value for an instant atwhich a value of the actuating variable, especially the output value, ismade to take effect is determined by linear combination of at leastthree of the acquired measured current values. This has the advantagethat a very simple and rapid extrapolation is able to take place.

In example embodiments, a model value for the angular position isdetermined from the measured electrical quantities of motor voltage andmotor current, the synchronous machine in particular not being equippedwith a sensor for the angular position. This has the advantage ofallowing a sensorless control for a synchronous machine.

In example embodiments, the effective inductivity has different valuesat different angular positions of the rotor. This, in particular, allowsthe rotor angle position to be detected even at very low rotationalspeeds and at standstill.

In example embodiments, the period length of the superimposed signalamounts to n-times the pulse-width-modulation time, n being an integralnumber that is greater than two. This has the advantage that thefundamental wave component is able to be separated from thehigh-frequency component, especially at an excellent signal-noise ratio.

In example embodiments, the value of the motor voltage to be set isdetermined following a specific time period Ttrl, the time period inparticular amounting to four times the pulse-width-modulation period, orto a higher integral multiple thereof. It is advantageous that ameasured current value is able to be determined in synchrony with thepulse-width modulation period, and that, following the determination offour measured current values, the fundamental wave component and thehigher-frequency component are able to be separated, the adaptationfunctions providing an excellent approximation of the determined currentcharacteristic.

In example embodiments the motor current is acquired synchronously withthe instants of the greatest change in the signal voltage. This offersthe advantage that a local extreme value is acquired as measured currentvalue in every instance. In this way the amplitude of thehigher-frequency current component is able to be determined veryprecisely.

In example embodiments, the current component induced by thesuperimposed voltage signal, i.e., the higher-frequency currentcomponent, is subtracted from the acquired current, and the motorcurrent determined in this manner is supplied, as actual value, to thefield-oriented control. This has the advantage of separating thesuperimpositioning, so that the fundamental wave component that affectsthe control response of the machine is utilized for the control. Thehigher-frequency current component is used for determining the angularposition of the rotor.

In example embodiments, an estimated flux vector is produced, whosedirection corresponds to the estimated rotor angle position, thisestimated flux vector being supplied to the flux model as support,

the flux model also being supplied with the individual value of the setoutput voltage and the acquired motor current. The estimated flux vectoris able to be regulated toward the flux vector determined from theoutput voltage and the motor current. This has the advantage that thehigh dynamic response of the flux model is able to be utilized also atvery low rotational speeds.

In example embodiments, a rotor flux vector is determined in the fluxmodel, and its difference in relation to the estimated rotor flux vectoris forwarded to a PI element,

the output signal of the PI element being used to reduce drift of anintegration element of the flux model. This has the advantage ofallowing a sensorless control, in which drift of the integration elementis able to be counteracted in such a way that the estimated flux vectoris regulated in the direction of the physical flux vector, so that, dueto the support of the flux model, no phase shift of the model fluxvector arises in relation to the actual flux vector.

In example embodiments, below a first rotational speed, the flux modelis supported by the rotor flux vector which is based on the estimatedrotor position angle, the amount, in particular, being specifiable, andthe direction being estimated from the current component induced by thesuperimposed voltage signal. This has the advantage that the flux modelmay even be used at very low rotational speeds and at standstill.

In example embodiments, above a second rotational speed, the flux modelis supported by a rotor flux vector whose amount is specifiable andwhose direction is perpendicular to the direction of the induced voltagevector formed on the input side of the flux model. This has theadvantage that a high induced voltage is generated above the secondrotational speed, so that an excellent position detection of the rotorangle is possible. Furthermore, the superimposed high-frequency voltageis able to be switched off in the upper rotational speed range, whichallows complete utilization of the maximum rectifier output voltage.

Between the first and the second rotational speeds, in exampleembodiments, the flux model is supported by a rotor flux vector which isformed as the sum from a first component of the rotor flux vector, basedon the estimated rotor position angle, and from another component of therotor flux vector whose amount is specifiable and whose direction isperpendicular to the direction of the induced voltage vector formed onthe input side of the flux model, the ratio of the components being afunction of the frequency. This has the advantage that in thistransitional rotational speed range, i.e., between the first and secondrotational speeds, a continual change between the feedback variablestakes place, so that fluctuations in the control response are avoided.

The first rotational speed, in particular, is smaller than the secondrotational speed.

Example embodiments of the present invention are explained in greaterdetail with reference to the appended Figures.

DETAILED DESCRIPTION

According to example embodiments of the present invention, an electricmotor, in particular a synchronous motor, is supplied by an inverter,which includes a control circuit in its signal electronics. Thesynchronous motor has a rotor fitted with permanent magnets. As analternative, however, it is also possible to use a rotor having afield-excitation winding, in which case the excitation winding will beinductively coupled or must be supplied with a unipolar current with theaid of brushes.

This control circuit, which therefore operates without sensor, i.e.,without an angle sensor, is supplied with the motor current as actualvalue.

The control circuit sets the motor voltage as output quantity byoutputting a particular pulse-width modulation ratio, which, whenaveraged across pulse-width-modulation period TPWM, results in the valueto be set. In each time interval Tctrl, the control circuit calculatesnew values for the actuating variables and thereby determines aparticular next output voltage value, i.e., a particular next value forthe motor voltage. This time period Tctrl amounts to fourpulse-width-modulation periods T_(PWM) in the exemplary embodimentshown.

To detect the angular position of the rotor, a high-frequency carriersignal is superimposed on the output variable of the control circuit. Inthe following text, the output variable is also referred to asfundamental wave, on which the carrier signal is superimposed.

The carrier signal is superimposed on the fundamental wave signaldetermined by the control, i.e., the motor voltage vector, in anestimated direction of the rotor. In this context, the motor voltagevector is representable as a vector which is rotatable in atwo-dimensional plane. The amplitude of the carrier signal isrepresentable in scalar manner, and the direction corresponds to theestimated direction of the rotor. The voltage vector to be supplied bythe inverter on the output side for the supply of the synchronous motorthus constitutes superimpositioning of the vector of the fundamentalwave and the carrier signal.

FIG. 1 illustrates idealized current and voltage characteristics by wayof example.

An exemplary setpoint voltage characteristic uSfx,y is shown at theinput terminals of the electric motor, in particular as manipulatedvariable of the current controller, and the superimposed high-frequencycarrier signal has been omitted. Voltage characteristic ucx of thehigh-frequency carrier signal has a period length Tc which exemplarilycorresponds to twice the length of pulse-width modulation period TPWM.

Because of the motor inductivity, the current characteristics associatedwith the voltage characteristics, i.e., current characteristic iScx,y ofthe high-frequency carrier signal, and current characteristic iSfx,yresulting from voltage characteristic uSfx,y, come about in a statorwinding of the electric motor. Resulting total current characteristiciSx,y in the stator winding of the electric motor is shown as well.

According to example embodiments of the present invention, the motorcurrent, i.e., the motor current vector, is acquired at the instantsshown in FIG. 1, that is to say, at sampling instant t0, samplinginstant t1, sampling instant t2, and sampling instant t3 of the currentacquisition.

The sampling instants are synchronous with the pulse-width modulationsignal. In addition, the sampling instants are also set in synchronywith the particular instants at which the rectangular carrier signalexhibits its jump positions. At these instants, the voltagecharacteristic of the carrier signal thus exhibits the greatest changesin voltage. In this manner the particular maxima and minima of thecurrent characteristic thus lie at the sampling instants, so that thecurrent is acquired at these extreme values in each case.

Using synchronous sampling, it is possible to determine the mentionedlocal maxima and minima of the current characteristic induced by thevoltage characteristic of the carrier signal. In so doing, a linearcurrent characteristic is assumed within a pulse-width modulationperiod, which constitutes an excellent approximation at a pulse-widthmodulation frequency of several kHz, such as approximately 4 kHz, 8 kHz,or even 16 kHz.

In contrast to using a bandpass filter for separating the high-frequencycarrier signal, no phase shift is therefore induced. This is so becausethe synchronous sampling according to the present invention allows adirect determination of the high-frequency current characteristic.

For the separation of the high-frequency carrier signal component fromthe fundamental wave component, it is assumed that the fundamental waveis a polynomial of the second order as a function of time, thispolynomial being defined by three parameters; it is also assumed thatthe carrier signal is composed of linear function segments, the carriersignal being in synchrony with the pulse-width modulation frequency. Theamplitude of the carrier signal and also the three parameters of thepolynomial are therefore able to be determined via the four acquiredcurrent values.

On the one hand, this provides an excellent approximation of the carriersignal characteristic and the fundamental wave characteristic and, onthe other hand, an extrapolation of the current value to be expected atinstant t4 is implemented. Instant t4 once again is evenly spaced apart,similar to sampling instants t0, t1, t2, t3. The instants thus have thesame time interval in relation to the preceding instant.

As mentioned earlier already, the values of the actuating variables,especially the motor voltage, are determined anew in each time intervalTctrl. The required computational steps are performed between instantst3 and t4. The input variable for the determination is the current valueof the fundamental wave component extrapolated for instant t4. In thisway, the actually existing current value at this instant is known veryprecisely, and an actual current value, without dead time, is utilizedfor determining the values of the actuating variables.

The high-frequency component is used to determine the rotor angle. Sinceit is not routed through a bandpass filter or similar means, it is notaffected by phase shift and allows a highly precise determination of therotor angle position. In particular, the high-frequency currentcomponent demodulated in this manner is not falsified even in thepresence of dynamic changes of the fundamental wave component. Thisapplies also when the employed motor exhibits significant saturationbehavior, as a result of which the current rises of the fundamental wavetake place in non-linear manner.

FIG. 2 shows the part of the control structure used for low rotationalspeeds of the rotor for one exemplary embodiment. Here, the supplyvoltage of the three-phase motor is supplied by means of output stage 1.The acquired motor current, i.e., stator current vector i_(S), isrepresented by the coordinate transformation T(x) in a coordinatesystem, which is rotated in relation to the stationary coordinate systemby means of the estimated rotor position angle. The stator currentvector is then forwarded to a decoupling module, which, as previouslydescribed, filters the high-frequency signal component from the acquiredmeasured current values and determines an extrapolated current valuei_(Sf4) for the fundamental wave component, which is forwarded to thecurrent controller, which includes PI element 3. Carrier signal voltageU_(C) is superimposed to the output signal of PI element 3, and theresult is supplied,as actuating variable, to output stage 1 operated inpulse-width-modulated manner.

The high-frequency signal component filtered out is {tilde over (θ)}_(e)transformed into a phase error, from which a reference angle for rotorposition θ_(e) is determined using estimated rotor position {tilde over(θ)}_(rc) angle. The model value for the rotor position thus determinedis used as direction for an estimated rotor flux vector, which isforwarded to a flux model 4. The system deviation of the rotor fluxvector is supplied to a PI element 5.

In the process, set voltage vector uS is reduced by the Ohmic voltagedrop RS×iS and then added, i.e., integrated, to the value of theestimated stator flux vector. From this, a flux component correspondingto the stator current is deduced, so that an estimated value for therotor flux vector is determined in this manner.

The argument of the estimated rotor flux vector is used for determiningthe reference angle for the rotor position {tilde over (θ)}_(re),together with ascertained phase error {tilde over (θ)}_(e).

The exemplary embodiment of FIG. 2 is advantageously able to be used forlow rotational speeds of the rotor.

FIG. 3 shows another exemplary embodiment, which is advantageously ableto be used for high and low frequencies.

In the process, the flux model for low rotational speeds is based on areference flux vector, whose direction was determined from thehigh-frequency demodulated carrier current, as described in connectionwith FIG. 2.

A rotor flux vector is likewise used as support of the flux model atvery high rotational speeds, its angular position, however, beingdetermined without use of the high-frequency carrier current. This angleis determined in such a way that it is directed perpendicular to thedirection of the induced voltage vector, which is determined on theinput side in the flux model. As a result, the feedback variable isimplemented such that, in the presence of vanishing drift of theintegration result of the integrating element in the flux model, thefeedback variable vanishes as well.

At frequencies in a specifiable transitional frequency range, which liesbetween the mentioned high and low rotational speeds, a mixture, inparticular linear superimpositioning as a function of the frequency, ofthe two rotor flux vectors is determined and then used as support of theflux model.

Additional details and features of example embodiments of the presentinvention are shown in the figures.

LIST OF REFERENCE CHARACTERS

1 output stage

2 decoupling module

3 PI element

4 flux model

5 PI element

U_(cx) voltage characteristic of the high-frequency carrier signal

i_(Scx,y) current characteristic of the high-frequency carrier signal

u_(Sfx,y) current characteristic of the fundamental wave component ofthe motor voltage, in particular as manipulated variable of the currentcontroller

i_(Sfx,y) current characteristic resulting from U_(Sfx,y) in a statorwinding of the electric motor

i_(Sx,y) resulting total current characteristic in a stator winding ofthe electric motor

T_(ctrl) period duration of the current controller

T_(c) period duration of the high-frequency carrier signal

T_(PWM) length of the pulse-width modulation period

t₀ sampling instant of the current acquisition

t₁ sampling instant of the current acquisition

t₂ sampling instant of the current acquisition

t₃ sampling instant of the current acquisition

t₄ prediction instant for the current

PMSM synchronous motor having permanent magnets disposed on its rotor

1-15. (canceled)
 16. A method for determining a rotor position of asynchronous machine operated in field-oriented manner, which has aneffective inductivity that is dependent on the rotor position,comprising: acquiring a motor current; setting a motor voltage inaccordance with a pulse-width-modulation method; superimposing a signalwhich is synchronous with a pulse-width-modulation frequency on themotor voltage value to be set; acquiring values of the motor current insynchrony with the pulse-width modulation frequency; determining acurrent component induced by the superimposed voltage signal and atleast one of (a) a residual current component and (b) a fundamental wavecomponent; using the current component induced by the superimposedvoltage signal to determine an estimated rotor angle position, whosephase error in relation to an actual rotor angle position is reduced bya flux model; and supplying the residual current component to a currentcontroller.
 17. The method according to claim 16, wherein a number ofmeasured motor current values are used for determining parametersprovided in a first parameter-based function assigned to the fundamentalwave component, and in a second parameter-based function assigned to thecurrent component induced by the superimposed voltage signal, a sum ofparameter-based functions at least one of (a) being adapted to theacquired current values and (b) being adapted as fitting functions. 18.The method according to claim 17, wherein at least one of (a) the firstparameter-based function is a polynomial of the n-th order, a number ofmeasured motor current values being greater than n by one, and (b) thesecond parameter-based function is at least one of (i) a triangularsignal and (ii) a function composed of linear segments, its amplitudebeing a parameter and its frequency being synchronous with thepulse-width modulation frequency.
 19. The method according to claim 18,wherein the number is four or greater than four.
 20. The methodaccording to claim 17, wherein following the determination of theparameters of the parameter-based functions, a current value isextrapolated for an instant at which the values of at least one of (a)an actuating variable and (b) an output voltage are made to take effect,so that a synchronous current value is used for determining the valuesof the actuating variable.
 21. The method according to claim 16, whereinan amplitude of the current component induced by the superimposedvoltage signal is determined from a linear combination of at least threeof the acquired measured current values.
 22. The method according toclaim 20, wherein an extrapolated current value for an instant at whicha value of at least one of (a) the actuating variable and (b) an outputvoltage is made to take effect is determined by a linear combination ofat least three of the acquired measured current values.
 23. The methodaccording to claim 16, wherein a model value for the angular position isdetermined from measured electrical quantities of motor voltage andmotor current, the synchronous machine not being equipped with anangular position sensor.
 24. The method according to claim 16, whereinat least one of (a) an effective inductivity has different values atdifferent rotor angle positions and (b) a period length of thesuperimposed signal amounts to n times the pulse-width modulationperiod, n being an integral number that is greater than one.
 25. Themethod according to claim 16, wherein at least one of (a) the motorvoltage value to be set is determined following an individual timeperiod amounting to four times the pulse-width-modulation period or to ahigher integral multiple thereof and (b) the motor current is acquiredsynchronously with instants of greatest change in the signal voltage.26. The method according to claim 16, wherein the current componentinduced by the superimposed voltage signal is subtracted from theacquired current, and the motor current determined in this manner isforwarded, as an actual value, to a field-oriented control.
 27. Themethod according to claim 16, wherein an estimated flux vector isproduced, whose direction corresponds to the estimated rotor angleposition, and the estimated flux vector is supplied to the flux model assupport, the flux model also being supplied with an individual value ofthe set output voltage and the acquired motor current.
 28. The methodaccording to claim 27, wherein a rotor flux vector is determined in theflux model, whose difference in relation to the estimated rotor fluxvector is forwarded to a PI element, an output signal of the PI elementbeing used to reduce drift of an integration element of the flux model.29. The method according to claim 16, wherein at least one of (a) belowa first rotational speed, the flux model is supported by the rotor fluxvector based on the estimated rotor position angle, the amount beingspecifiable, and a direction being estimated from the current componentinduced by the superimposed voltage signal, (b) above a secondrotational speed, the flux model is supported by a rotor flux vectorwhose amount is specifiable and whose direction is perpendicular to thedirection of the induced voltage vector formed on the input side of theflux model, (c) between the first and the second rotational speeds, theflux model is supported by a rotor flux vector formed as the sum from afirst component of the rotor flux vector based on the estimated rotorposition angle, and from another component of the particular rotor fluxvector whose amount is specifiable and whose direction is perpendicularto the direction of the induced voltage vector formed on the input sideof the flux model, the ratio of the components being a function of thefrequency.
 30. The method according to claim 29, wherein the firstrotational speed is smaller than the second rotational speed.